Timing controlled ac to dc converter and method

ABSTRACT

A timing controlled converter and method for converting a time varying input signal to a regulated DC output voltage for application to a load circuit. A feedback loop is employed as a control means for switchably coupling the time varying input signal to the load circuit for controlled periods of time in a manner so as to provide an average load voltage equal to a reference voltage. The duration of the controlled periods of time is a function of: the difference between the time varying input signal and the output voltage; and the integral of the difference between the output voltage and the reference voltage.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application is a continuation of U.S. patent application Ser. No. 12/963,598 filed on Dec. 8, 2010, which issued on Jan. 20, 2015 as U.S. Pat. No. 8,937,825 and claims the benefit under 35 U.S.C. §119(e) to U.S. Provisional Application No. 61/285,520, filed on Dec. 10, 2009, both of which are incorporated herein by reference in their entirety.

FIELD OF THE INVENTION

This disclosure relates generally to electronic integrated circuits for use typically in implantable biomedical devices. More specifically, the disclosure relates to a timing controlled AC to DC converter.

BACKGROUND OF THE INVENTION

Many biomedical implantable devices are powered from an external time varying magnetic source. The magnetic source is inductively coupled to a coil housed within the implantable device to induce an AC voltage in the coil which is then converted or rectified to a regulated DC voltage for use to power device electronics. As a general matter, higher supply voltages are often required for analog circuits, such as stimulation circuits in implantable neuro-prosthetic devices, whereas lower supply voltages are usually required for digital circuits and processors. Accordingly, a rectified or regulated DC voltage is typically maintained at a higher value for analog circuits and a linear regulator is implemented to convert the higher value DC voltage to a lower supply voltage for the digital circuits. An example of an implantable stimulator that includes both analog and digital circuitry and voltage supplies providing a range of output voltages is found in U.S. Pat. No. 6,185,452 to Schulman, et al. As has been recognized, the approach taken in the art to generate multiple supply voltages as described above, is not very power efficient especially when considering the limited power availability from weak inductive coupling of the implanted device with the external time varying magnetic source. Attempts have been made to improve power efficiency such as the use of Buck converters and switchable capacitor converters; however they may not be suitable for neuro-prosthetic and biomedical implant applications due to the limited space available within the devices which typically can only accommodate a few discrete components. Accordingly, what is needed to satisfy the shortcomings of the prior art is an approach based on a direct conversion of the induced AC voltage to a regulated DC voltage which achieves high conversion efficiency using circuitry sufficiently small to be housed in small implantable devices.

SUMMARY

An example embodiment of the invention discloses a timing controlled AC to DC converter which supplies a regulated output voltage to a load circuit. The converter switchably couples a time varying input signal for a controlled period of time directly to the load circuit in a manner such that the average value of the regulated output voltage equals a preselected reference voltage. An embodiment of the invention comprises an integrator arranged to integrate the difference between the output or load voltage and the preselected reference voltage to thereby generate a threshold or control signal. A switch is arranged to switchably couple the time varying input signal to the load circuit for controlled periods of time. A controlled period of time commences when the time varying input signal becomes greater than the output voltage and terminates when the time varying input signal becomes greater than the value of the control signal. The duration of the controlled periods of time, so defined, causes the converter to provide an average value of the output voltage being equal to the preselected reference voltage while minimizing power loss and significantly improving converter efficiency.

BRIEF DESCRIPTION OF DRAWINGS

The accompanying drawings, which are incorporated into and constitute a part of this specification, illustrate one or more embodiments of the present disclosure, and together with the description of example embodiments, serve to explain the principles and implementations of the disclosure.

FIGS. 1A-1C show circuit implementations of conventional power management units known in the art. More particularly, FIG. 1A shows an implementation of a power management unit using a switched-capacitor (SC) converter; FIG. 1B shows an implementation of a power management unit using a buck converter; and FIG. 1C shows an implementation of a power management unit using a linear regulator.

FIG. 2 shows an implementation of a timing controlled converter in accordance with an embodiment of the present invention.

FIG. 3 shows a voltage waveform that illustrates the controlled period of time during which the converter couples the time varying input signal to the load circuit.

FIG. 4 shows a block diagram of an embodiment of the timing controlled converter of the present invention.

FIG. 5 shows a detailed block diagram of the converter of FIG. 4.

FIG. 6 shows voltage waveforms related to the operation of the timing controlled converter of FIG. 5.

FIGS. 7A and 7B show first and second comparators, respectively, used in an implementation of the timing controlled converter of FIG. 4.

FIG. 8 shows an implementation of a boosted switch driver that is used in an implementation of the timing controlled converter of FIG. 4.

DETAILED DESCRIPTION

Power management units or voltage converter circuits housed within implantable medical devices are generally used to generate supply voltages required for operation of electronic circuits within the devices. As an illustration, FIGS. 1A-1C show conventional implementations of power management units housed within implantable devices. Power management units typically generate a high voltage supply V_(C) (100), which may be used by a stimulator to generate stimulation current pulses and a low output voltage V_(DD) (102), which may be used for powering various low voltage circuits. According to several embodiments of the present invention, converter circuits are assumed to receive power through inductive coupling from a radio frequency (RF) magnetic field, which is the case for many implantable biomedical devices. A metric used to characterize the converter is power efficiency, denoted as q. Power efficiency η is defined as a ratio between the output power of the converter to the input power supplied to the converter. Consequently, a high power efficiency η signifies that a large proportion of the input power is available as output power, which is generally desired.

Specifically, according to several embodiments of the present invention converters in implantable biomedical devices are powered from an external time varying magnetic source (not shown). The external magnetic source is magnetically coupled to a coil L_(E) (115). This magnetic coupling results in an induced voltage V_(RF) (110) across the coil L_(E) (115). A tuning capacitor C_(E) (120) is designed to resonate at an operating frequency f_(in). A conventional rectifier (125) is used to generate the high supply voltage V_(C) (100) measured between Vc and ground (105). The high supply voltage V_(C) (100) derives from the positive peak value and a negative peak value of the induced voltage V_(RF) (110).

In the present disclosure, the induced voltage V_(RF) (110) across the coil L_(E) (115) is assumed to be periodic and has a peak value higher than the required high supply voltage V_(C) (100). In particular, the induced voltage V_(RF) (110) is arbitrarily assumed to be a sinusoidal wave. Consequently, after rectifying the induced voltage V_(RF) (110), the rectifier (125) can produce the high supply voltage V_(C) (100) equal to or less than the peak voltage of the induced voltage V_(RF) (110).

The low supply voltage V_(DD) (102), on the other hand, can be generated by taking the high supply voltage V_(C) (100) of the rectifier (125) and converting it to a desired value for the low supply voltage V_(DD) (102). Specifically, DC to DC converters, such as switched-capacitor converters, buck converters, and linear regulators, may be used to generate the low supply voltage V_(DD) (102).

As seen from FIGS. 1A-1C (and later in FIG. 2), the coil L_(E) (115), tuning capacitor C_(E) (120), and rectifier (125) are common to the different implementations of the converters. Additionally, the induced voltage V_(RF) (110), high supply voltage V_(C) (100), and low supply voltage V_(DD) (102) refer to the same voltages in each of FIGS. 1A-1C. Since the components (115, 120, 125) and the voltages (110, 100, 102) referred to are the same for different implementations of converter units, the same reference numerals are used to refer to the same components (115, 120, 125) and voltages (110, 100, 102) in each of FIGS. 1A-1C. Additionally, a load circuit, which is not part of the converter unit, comprises a bypass capacitor C_(B) (130) and a load resistor R_(L) (135). The bypass capacitor C_(B) (130) is used to remove any AC voltage components from the low supply voltage V_(DD) (102) such that only a DC voltage component is applied to the load resistor R_(L) (135). The same off-chip load circuit will be used in many subsequent figures.

FIG. 1A shows an implementation that utilizes a switched-capacitor (SC) converter (140) for generating the low supply voltage V_(DD) (102). The SC converter (140) can achieve high power efficiency η. However, the SC converter (140) requires a plurality of discrete capacitors (145) comprising C₁ through C_(N).

The number of discrete capacitors (145) in a typical design, denoted as N, is approximately given by the equation N=V_(C)/V_(DD)−1. Consequently, for a typical high supply voltage V_(C) (100) of 10 V to 15 V and typical low supply voltage V_(DD) (102) of 3 V, the number N of discrete capacitors (145) ranges from 2 to 4. This number N does not include the bypass capacitor C_(B) (130).

FIG. 1B shows an implementation that utilizes a buck converter (150) for generating the low supply voltage V_(DD) (102). The number of discrete components is reduced in this implementation because of using a buck converter. However, the disadvantage of this implementation is the necessity for the inclusion of an inductor L_(B) (155).

FIG. 1C shows an implementation of a converter unit that utilizes a linear regulator (160) for generating the low supply voltage V_(DD) (102). This implementation that utilizes the linear regulator (160) does not require extra discrete components aside from the bypass capacitor C_(B) (130). However, current flowing to the load resistor R_(L) (135) also flows through the linear regulator (160). As a result, an equivalent power of (V_(C)−V_(DD))·V_(DD)/R_(L) is dissipated by the linear regulator (160) for a total power consumption of V_(C)−V_(DD)/R_(L) drawn from the high supply voltage V_(C) (100).

With continued reference to FIG. 1C, for a value of 3V for the low output voltage V_(DD) (102) and a range of 10V to 15V for the large output voltage V_(C) (100), power efficiency η is between 20% and 30%, which is not very efficient. Although the implementations using the SC converter (140) and the buck converter (150) generally have higher power efficiencies compared to the implementation using linear regulators, especially for high load conditions, the implementation of the converter unit using the linear regulator (160), shown in FIG. 1C, is often the only viable choice for implantable devices due to their small size.

FIG. 2 shows an implementation of an AC to DC converter of the present invention using a timing controlled rectifier (200), henceforth referred to as TCR. By comparing FIG. 2 with FIGS. 1A-1C, the coil L_(E) (115), tuning capacitor C_(E) (120), and rectifier (125) are common to the different implementations. The induced voltage V_(RF) (110), high supply voltage V_(C) (100), and low supply voltage V_(DD) (102) in FIG. 2 refer to the same voltages in each of FIGS. 1A-1C. The low supply voltage V_(DD) (102) may be considered the output voltage of the AC to DC converter circuit of the present invention. Additionally, the same off-chip load circuit comprising the bypass capacitor C_(B) (130) and the load resistor (135) is also used in FIG. 2.

The TCR (200) is adapted to generate the output voltage V_(DD) (102) directly from the induced voltage V_(RF) (110). The TCR (200) can achieve relatively good power efficiency without using extra discrete components aside from the bypass capacitor C_(B) (130).

The induced voltage V_(RF) (110) is directly applied to the rectifier (125) and to the TCR (200). The TCR (200) converts the induced voltage V_(RF) (110), which is an AC signal, into the required output voltage V_(DD) (102), which is a DC voltage. The rectifier (125) is used to generate the high supply voltage V_(C) (100) by rectifying the positive peak value and negative peak values of the induced voltage V_(RF) (110).

FIG. 3 shows an exemplary voltage waveform for the induced voltage V_(RF) (110) and also illustrates the method of setting the controlled period of time for coupling V_(RF) (110) to a load circuit, which in this case comprises bypass capacitor C_(B) (130) and load resistor R_(L) (135) and which also defines the basic operation of the TCR (200). As illustrated in FIG. 3, the controlled amount of time for coupling V_(RF) (110) to a load circuit is shown by a turn-on period Δt (300).

The TCR's (200) turn-on period Δt (300) occurs for a time period where the induced voltage V_(RF) (110) is slightly above the output voltage V_(DD) (102), as illustrated in FIG. 3. In particular, the TCR's (200) turn-on period Δt (300) is given as a difference between a turn-on time t_(s) (305) and a turn-off time t_(o) (310). During the turn-on period Δt (300), the TCR (200) conducts current from the voltage source V_(RF) (110) to the load circuit (capacitor C_(B) (130) and load resistor R_(L) (135)) otherwise identified as output voltage V_(DD) (102) such that the current is drawn from V_(RF) (110) and stored as accumulated charge in the bypass capacitor C_(B) (130). The turn-on time t_(s) (305) and the turn-off time t_(o) (310) will be described in more detail in relation to FIGS. 4 and 5, in which the operation of the TCR (200) will be described in more detail.

FIG. 4 shows an exemplary embodiment of the TCR (200). The same off-chip components, which comprise the bypass capacitor C_(B) (130) and load resistor R_(L) (135), as shown in FIG. 2 are also shown in FIG. 4. As shown in FIG. 4, the value of the output voltage V_(DD) (102) is set by a negative feedback control loop (hereinafter referred to as a “feedback loop”) that switchably controls the operation of transistor M_(SW) (405). Specifically, the feedback loop comprises a digital control unit (410), an integrator (415), a first comparator (420), a second comparator (430), a switch driver (445), and the transistor M_(SW) (405).

As a whole, the feedback loop is used to ensure that the transistor M_(SW) (405) turns on only during the portion of time that the induced voltage V_(RF) (110) is increasing in value. Specifically, with reference to the waveform on FIG. 3, the transistor M_(SW) (405) turns on only during the turn-on period Δt (300). When the induced voltage V_(RF) (110) becomes larger than the output voltage V_(DD) (102), the transistor M_(SW) (405) will turn on at the time t_(s) (305).

Specifically, the digital control unit (410), which comprises a specific arrangement of logic gates, takes a first comparator output V_(CO1) (435) and a second comparator outputs V_(CO2) (440) as its inputs and provides a first control voltage V_(on) _(—) _(b) (455) and a second control voltage V_(on) (460) as its outputs. Both V_(on) _(—) _(b) (455) and V_(on) (460) are applied as inputs to the switch driver (445). The switch driver (445) provides a switch driver voltage V_(BG) (465), which is applied to the gate of the transistor M_(SW) (405). Logic gates in the digital control unit (410) are designed in such a way as to ensure that transistor M_(SW) (405) turns on only during a portion of time that the induced voltage V_(RF) (110) is increasing in value.

The integrator (415) takes as its input the output voltage V_(DD) (102) and a reference voltage V_(ref) (400). The integrator (415) integrates the difference between the output voltage V_(DD) (102) and the reference voltage V_(ref) (400) and provides thereby a resultant integrated value identified as the integrator output voltage V_(int) (425). The voltage V_(int) (425) is used as a threshold voltage or control signal for application to the second comparator CO₂ (430).

The first comparator CO₁ (420) has as its inputs the induced voltage V_(RF) (110) and the output voltage V_(DD) (102). The first comparator CO₁ (420) detects when the induced voltage V_(RF) (110) becomes larger than the output voltage V_(DD) (102). When the induced voltage V_(RF) (110) becomes larger than the output voltage V_(DD) (102), the first comparator output voltage V_(CO1) (435) output from the first comparator CO₁ (420) will cause transistor M_(SW) (405) to turn on at the turn-on time t_(s) (305). More specifically, the voltage M_(CO1) (435) is applied to the digital control unit (410), which outputs a first output V_(on) _(—) _(b) (455) and a second output V_(on) (460) to the switch driver (445). The switch driver (445) provides a switch driver output V_(BG) (465) to the gate of the transistor M_(SW) (405), which turns on the transistor M_(SW) (405). When the transistor M_(SW) (405) is turned on, a load current I_(L) (445) is supplied from V_(RF) (110) to the load resistor R_(L) (135).

The second comparator CO₂ (430) has as its inputs the voltage V_(RF) (110) and the integrator output voltage V_(int) (425). The second comparator CO₂ (430) detects when the voltage V_(RF) (110) becomes greater than the integrator output voltage V_(int) (425). When V_(RF) (110) becomes greater than the voltage V_(int) (425), output voltage V_(CO2) (440) of the second comparator CO₂ (430) will cause transistor M_(SW) (405) to turn off at the turn-off time t_(o) (310). The truth table (Table 1) shown below illustrates the relationship between the identified voltages and the corresponding state of transistor M_(SW) (405).

TABLE 1 Voltage Condition V_(BG1) Transistor M_(SW) V_(CO1) High if V_(RFa) > V_(DD) High ON Low if V_(RFa) < V_(DD) Low OFF V_(CO2) High if V_(RFa) > V_(int) High OFF Low if V_(RFa) < V_(int) Low ON

For the TCR (200) shown in FIG. 4, a load current I_(L) (450) flows from voltage source V_(RF) (110) to the load circuit. Operation of the feedback loop, and thus operation of the TCR (200) itself, is as follows. When the average load current I_(L) (450) increases, the average output voltage V_(DD) (102) decreases. Since the integrator output voltage V_(int) (415) is a function of the difference between the voltages V_(DD) (102) and V_(ref) (400), the integrator output voltage V_(int) (415) increases when V_(DD) (102) decreases. Since the first comparator output V_(CO1) (435) is a function of the difference between the voltages V_(RF) (110) and V_(DD) (102), the first comparator CO₁ (420) will have a logic high at the output earlier when the voltage V_(DD) (102) decreases, which causes transistor M_(SW) (405) to turn on earlier. Since the second comparator output V_(CO2) (440) is a function of the difference between the voltages V_(RF) (110) and V_(int) (415), the second comparator CO₂ (430) will have a logic high at the output later when the voltage V_(int) (415) increases, which would cause transistor M_(SW) (405) to turn off later. Hence, the turn-on period Δt (300) for the transistor M_(SW) (405) will become longer and more charge will be accumulated in the bypass capacitor C_(B) (130).

Steady state is reached when an average value of the output voltage V_(DD) (102) becomes equal to the reference voltage V_(ref) (400). In the steady state, voltage V_(int) (425) will become nearly constant and the transistor M_(SW) (405) will turn on and off for an amount of time needed to accumulate just enough charge to keep the average value of the output voltage V_(DD) (102) equal to the reference voltage V_(ref) (400). As a result, the average value for the output voltage V_(DD) (102) is regulated to have a value equal to the reference voltage V_(ref) (400).

With reference to FIG. 4 and later to FIG. 6, it should be noted that transistor M_(SW) (405) and M_(SW1) (563) in the case of FIG. 6, will be turned on when the polarity of V_(RF) and V_(DD) (535) are the same. Furthermore, it should also be noted that although FIG. 3 and FIG. 6, for example, show and describe V_(DD) as having a positive polarity, the invention contemplates a configuration for operation with V_(DD) having a negative potential with respect to ground. Thus in order to avoid any confusion regarding comparing voltages when referring to relative magnitudes with respect to the use of the terms “greater than”, “larger than”, “exceeds”, and the like, it is to be understood that absolute values of voltage magnitudes are to be considered.

It should be noted that due to any delays attributable to the comparators CO₁ (420) and CO₂ (430), the voltage V_(RF) (110) at the turn off time t_(o) (305), denoted as a voltage V_(OFF) (315) in FIG. 3, may be slightly higher than the voltage V_(int) (425). However, the feedback loop will adjust the turn-off time t_(o) (305) so as to maintain the average value of the output voltage V_(DD) (102) equal to the reference voltage V_(ref) (400).

Consider an ideal case where the transistor M_(SW) (405) has a turn-on resistance R_(SW) (not shown) that is nearly zero and where the comparator (420, 430) delays are zero. The turn-on period Δt (300) required to accumulate sufficient charge on the bypass capacitor C_(B) (130) will approach zero. In this ideal case, the voltage V_(OFF) (315) will equal the output voltage V_(DD) (102). Additionally, in this ideal case, no power is dissipated by the transistor M_(SW) (405) and the power efficiency η for converting power from the induced voltage V_(RF) (110) to the output voltage V_(DD) (102) will approach 100%.

As is recognized, transistor M_(SW) (405) has a finite turn-on resistance R_(SW), especially when a high-voltage MOSFET is used for the transistor M_(SW) (405). In such cases, the power efficiency η may be about 50% to 90% depending on the frequency and the amplitude of the induced voltage V_(RF) (110), the value of V_(DD), the turn-on period Δt and the resistance of the transistor M_(SW). As the frequency of the induced voltage V_(RF) (110) increases, the power efficiency η decreases due to higher dynamic power dissipation in driving M_(SW) more frequently. Compared to the implementation utilizing the linear regulator (160) as shown in FIG. 1C, it can be shown that in a case where V_(D)>V_(DD)/η, the embodiment of the converter using TCR (200) will provide some power savings by a percentage of [1−V_(DD)/(ηV_(C))] compared to the use of the linear regulator (160). For instance, given an exemplary situation where V_(C)=10 V, V_(DD)=3 V and n=50%, the TCR (200) will provide a power saving of 40% compared to the use of the linear regulator (160).

Referring now to FIG. 5 there is shown an embodiment of the converter that incorporates the timing controlled rectifier (200) shown in FIG. 4. A bypass capacitor C_(B) (580) and a load R_(load) (585) are off-chip components and thus are not part of the converter.

The converter unit in FIG. 5 is inductively powered from an external magnetic source (not shown). The external magnetic source induces a voltage V_(RF) (500) across receiving coil L_(E) (502). Capacitor C_(E) (504) and coil L_(E) (502) form a parallel resonant circuit and the value of tuning capacitor C_(E) (504) is selected to establish a circuit resonant frequency of f_(in). The induced voltage V_(RF) (500) comprises a first voltage rail RF_(A) (506) corresponding to a first voltage V_(RFa) (507) and a second voltage rail RF_(B) (508) corresponding to a second voltage V_(RFb) (509). The power management unit further comprises a full-wave rectifier (510), a first converter core (515), a second converter core (520), and integrator (525). The full-wave rectifier (510) generates a high supply voltage V_(C) (530), which is generally used to supply voltage to the converter cores (515, 520) in the power management unit as well as analog circuits in implantable devices (not shown).

The timing controlled rectifier (200) in FIG. 4 comprises one of the converter cores (515, 520) and the integrator (525) shown in FIG. 5. The timing controlled rectifier (200) generates an output voltage V_(DD) (535), which is generally used to power digital control circuits and communication circuits in implantable devices (not shown).

The first converter core (515) comprises a first comparator CO₁ (545), a second comparator CO₂ (550), a digital control unit (555), and a boosted switch driver (560). Note that the boosted switch driver (560), shown in FIG. 5, is equivalent to the combination of the switch driver (445) and the transistor M_(SW) (405), shown in FIG. 4. The second converter core (520) comprises components, not shown in FIG. 5, similar to those in the first converter core (515). Specifically, the second converter core (520) also comprises its own first comparator, second comparator, digital control unit, and boosted switch driver. An implementation of the integrator (525) is shown in FIG. 5 and comprises a transconductance amplifier (526), a capacitor C_(C) (527), and a resistor R_(C) (528).

As previously described with reference to FIG. 4, basic functionality of the converter cores (515, 520), the integrator (525), the digital control unit (555), and the boosted switch driver (560), which form a feedback loop, is to set the average value of the output voltage V_(DD) (535) equal to a constant reference voltage V_(ref) (540).

The integrator (525) has as one input the preselected reference voltage V_(ref) (540) and the output voltage V_(DD) (535) as another input and provides as an output, a voltage V_(int) (529) equal to the integral of the difference of V_(ref) (540) and V_(DD) (535). The first comparator CO₁ (545) has as one input V_(RFa) (507) and as another input V_(DD) (535) and provides as an output voltage V_(CO1) (547), which is a function of the difference between V_(RFa) (507) and V_(DD) (535). In a similar fashion, the second comparator CO₂ (550) has as inputs V_(RFa) (507) and V_(int) (529) and provides V_(CO2) (552) as its output. A more detailed description of the functionality of the converter cores (515, 520) and the integrator (525) is given with reference to both FIGS. 5 and 6.

As shown in FIG. 6, during a first time interval a (600), where the first voltage V_(RFa) (507) is increasing, a first transistor M₁ (512) has the voltage V_(RFa) (507) applied at its gate, the voltage V_(RFb) (509) applied at its drain, and ground tied to its source. Since the gate voltage is sufficiently large during the time interval a (600), the second voltage rail RF_(B) (508) is tied to the ground terminal. Consequently, the first transistor M₁ (512) is switched on and the second voltage rail RF_(B) (508) is tied to ground.

When the voltage V_(RFa) (507) becomes greater than the output voltage V_(DD) (535), the first comparator output voltage V_(CO1) (547) of the first comparator CO₁ (545) is set to high. When the voltage V_(CO1) (547) is high, the digital control unit (555) sets a gate drive voltage V_(BG1) (562) in the boosted switch driver (560) to high. When the gate drive voltage V_(BG1) (562) is high, a transistor M_(SW1) (563) is turned on, allowing current to flow from the first voltage rail RF_(A) (506) to the bypass capacitor C_(B) (580) and the load R_(load) (585). As the result of an equivalent finite impedance attributable to the power receiving coil L_(E) (502) and the tuning capacitor C_(E) (504), the voltage V_(RFa) (507) will be distorted as shown in FIG. 6. As previously mentioned, the feedback loop comprising the converter cores (515, 520), the integrator (525), the digital control unit (555), and the boosted switch driver (560) is utilized to set the average value of the output voltage V_(DD) (535) equal to the constant reference voltage V_(ref) (540).

The integrator (525) integrates the average difference between the constant reference voltage V_(ref) (540) and the output voltage V_(DD) (535). The integrator (525) outputs an integrator output voltage V_(int) (529), which serves as a threshold value for the second comparator CO₂ (550). When the voltage V_(RFa) (507) becomes greater than the voltage V_(int) (529), such as at point b (605), the second comparator output voltage V_(CO2) (552) is set to high. When the voltage V_(CO2) (552) is high, the digital control unit (555) sets the gate drive voltage V_(BG1) (562) to low and thus turns off the transistor M_(SW1) (563). Current flow will cease for a time interval c (610) so that the amount of current delivered to the load R_(load) (585) will keep the average value for the output voltage V_(DD) (535) equal to the constant reference voltage V_(ref) (540). Consequently, duration of time when transistor M_(SW1) (563) is turned off depends on a load current I_(L) (590).

To prevent current from flowing back from the output voltage V_(DD) (535) to the first voltage rail RF_(A) (506), transistor M_(SW1) (563) is turned off before the voltage V_(RFa) (507) reaches a peak value V_(P) (615), as shown in FIG. 6. Current flowing from the output voltage V_(DD) (535) to the first voltage rail V_(RFa) (506) would cause the converter to lose regulation and reduce power efficiency η. High values for power efficiency η are obtained if the transistor M_(SW1) (563) turns on when the voltage V_(RFa) (507) is nearly equal to the voltage V_(DD) (535).

One method to prevent current from flowing back from the output voltage V_(DD) (535) to the first voltage rail RF_(A) (506) requires design of the digital control unit (555) such that transistor M_(SW1) (563) will not turn on when the voltage V_(RFa) (507) decreases from its peak value V_(P) (615). As seen in FIG. 6, the bias voltage V_(BG1) (562) is set to low and thus transistor M_(SW1) (563), is turned off for the time that the voltage V_(RFa) (507) is decreasing from its peak value V_(P) (615).

It should be noted that in FIG. 5 the second converter core (520) is illustrated by a black box for increased clarity. According to several embodiments of the invention, the second converter core (520) comprises a first comparator, second comparator, digital control unit, and boosted switch driver that parallel those used to implement the first converter core (515). The components of the second converter core (520) are not shown in FIG. 6. When the induced voltage V_(RF) (500) enters a second half of its cycle, specifically the cycle where the voltage V_(RFb) (509) starts increasing, the first voltage rail RF_(A) (506) is switched to ground by the second transistor M₂ (513). Consequently, operation similar to that described for the time periods a (600) and c (610) will occur, except operation will involve the second converter core (520) as opposed to the first converter core (515).

FIGS. 7A and 7B show a possible implementation of the first comparator CO₁ (545) and the second comparator CO₂ (550), respectively. The implementations of the comparators (545, 550) are based on an exemplary common-source amplifier topology. FIG. 7A shows one possible implementation of the first comparator CO₁ (545) shown in FIG. 5. The voltages V_(RFa) (507), V_(DD) (535), and V_(CO2) (552) shown in FIG. 7B refer to the same voltages V_(RFa) (507), V_(DD) (535), and V_(CO2) (552) shown in FIG. 5 and are thus given the same reference numerals in both figures. The first comparator CO₁ (545) has the voltages V_(RFa) (507) and V_(DD) (535) as its inputs and provides the voltage V_(CO1) (547) as its output. The first comparator (545) compares the first voltage V_(RFa) (507) and the output voltage V_(DD) (535). In a first case where the voltage V_(RFa) (507) becomes greater than the voltage V_(DD) (535), a third transistor M₃ (700) turns on and comparator output voltage V_(CO1) (552) is high. In a second case where the voltage V_(RFa) (507) is less than the voltage V_(DD) (535), the third transistor M₃ (700) turns off and the comparator output voltage V_(CO1) (547) is low.

To maximize power efficiency q, the voltage V_(CO1) (547) should be high when the voltage V_(RFa) (507) is just slightly larger than the voltage V_(DD) (535). One method to achieve this is by minimizing delay attributable to the first comparator CO₁ (545). To minimize delays due to the first comparator CO₁ (545), a resistor R₁ (720) is used in comparator CO₁ (545). The resistor R₁ (720) introduces a small offset voltage V_(offset) (722). As a result of the small offset voltage V_(offset) (722), the comparator output voltage V_(CO1) (547) starts to turn high when the voltage V_(RFa) (507) is just slightly larger than the voltage V_(DD) (535) minus the offset voltage V_(offset) (737). The first comparator CO₁ (545) sets the voltage V_(CO1) (547) to high earlier than it would have set the voltage V_(CO1) (547) to high had there been no offset voltage V_(offset) (722). The earlier time at which the voltage V_(CO1) (547) is set to high compensates for any delay attributable to the first comparator CO₁ (545).

For a given bias voltage V_(bn) (725) set on a fourth transistor M₄ (705) and a sixth transistor M₆ (715), the resistance of the resistor R₁ (720) is optimized for a frequency of 1 MHz for f_(in). A capacitor C₁ (730) is used to minimize fluctuations on the gate of transistor M₃ due to coupling between the output voltage V_(DD) (535) and the voltage V_(RFa) (507) through a parasitic capacitance C_(GS3) (not shown) that exists between the gate and source of the third transistor M₃ (700).

FIG. 7B shows one possible implementation of the second comparator CO₂ (550) shown in FIG. 5. The voltages V_(RFa) (507), V_(int) (529), and V_(CO1) (547) shown in FIG. 7B refer to the same voltages V_(RFa) (507), V_(int) (529), and V_(CO1) (547) shown in FIG. 5 and are thus given the same reference numerals in both figures.

The second comparator CO₂ (550) has as inputs V_(RFa) (507) and V_(int) (529) and V_(CO2) (552) as its output. The second comparator CO₂ (550) compares the values of V_(RFa) (507) and V_(int) (529) and in a first case where the voltage V_(RFa) (507) exceeds the voltage V_(int) (529), transistor M₇ (750) turns on and comparator output voltage V_(CO2) (552) is high. Conversely, in a second case where V_(int) (529) exceeds V_(RFa) (507), transistor M₇ (750) turns off and comparator output voltage V_(CO2) (552) is low.

An exemplary source follower configuration, comprising transistor M₉ (760) and transistor M₁₀ (765), is used for its voltage buffering characteristic. Specifically, the source follower configuration buffers the integrator output voltage V_(int) (529) from anomalies due to coupling between the voltage V_(RFa) (507) and the voltage V_(int) (529) through a parasitic capacitance C_(GS7) (not shown) that exists between gate and source of transistor M₇ (750).

Any voltage offsets effecting V_(RFa) (507) and V_(int) (529) due to a source to gate voltage V_(SG7) (752) of M₇ (750) and a gate to source voltage V_(GS9) (762) of transistor M₉ (760) is automatically compensated for by way of the feedback loop. As noted earlier in relation to FIG. 5, the feedback loop comprises the integrator (525), comparators (545, 550), digital control unit (555), and boosted switch driver (560). The feedback loop will adjust the voltage V_(int) (529) to compensate for any comparator offsets due to the voltages V_(SG7) (752) and V_(GS9) (762). Since the voltage V_(int) (529) is the output of the integrator (525), as shown in FIG. 5, the value of the output voltage V_(DD) (535) is influenced by any offset voltages attributable to transconductance amplifier (526).

As previously mentioned with regard to FIGS. 5, 7A, and 7B, in order to obtain a high power efficiency η, the output voltage V_(DD) (535) is driven close to the voltage V_(RFa) (507) by the transistor M_(SW1) (563) and the gate drive voltage V_(BG1) (562). Similarly, the output voltage V_(DD) (535) is driven close to the voltage V_(RFb) (509) by a transistor M_(SW2) and a gate drive voltage V_(BG2). The transistor M_(SW2) and gate drive voltage V_(BG2) are not directly shown in FIG. 5. They are implicitly found in the second converter core (520), and the transistor M_(SW2) and gate drive voltage V_(BG2) parallel the transistor M_(SW1) (563) and the gate drive voltage V_(BG1) (562), respectively, of the first converter core (515). Hence, the transistors M_(SW1) (563) and M_(SW2) are generally designed to have low on-resistance which can be accomplished by increasing gate drive voltages V_(BG1) (562) and V_(BG2) (not shown) of the transistors M_(SW1) (563) and M_(SW2) (not shown), respectively.

FIG. 8 shows an implementation of the boosted switch driver (560) shown in FIG. 5. Specifically, the boosted switch driver (560) is used for generating a large value for the gate drive voltage V_(BG1) (562). The voltages V_(RFa) (507), V_(DD) (535), V_(C) (530), and V_(BG1) (562) shown in FIG. 8 refer to the same voltages V_(RFa) (507), V_(DD) (535), V_(C) (530), and V_(BG1) (562) shown in FIG. 5 and are thus given the same reference numerals in both figures. Similarly, the transistor M_(SW1) (563) shown in FIG. 8 refers to the same transistor M_(SW1) (563) shown in FIG. 5 and is thus given the same reference numeral in both figures.

When a driver input ‘IN’ (800) (same as the output of the digital control (555) shown in FIG. 5) is low, capacitor voltage V_(C2) (805) is driven to a high voltage by a first inverter U₁ (810) through capacitor C₂ (815). The voltage V_(C2) (805) is applied to the gate of transistor M₁₆ (840) and transistor M₁₇ (845), and the source of transistor M₁₈ (850). When the voltage V_(C2) (805) is high, transistor M₁₇ (845) is turned on due to the high gate voltage applied to transistor M₁₇ (845) and transistor M₁₈ (850) is turned off due to the high source voltage applied to transistor M₁₈ (850). As a result, the high supply voltage V_(C) (530) is applied to capacitor C₃ (855), charging capacitor C₃ (855) such that capacitor voltage V_(C3) (860) is close to the high supply voltage V_(C) (530). The gate drive voltage V_(BG1) (562) is driven to ground through transistor M₁₄ (830) and transistor M₁₃ (825).

When the driver input ‘IN’ (800) is high, capacitor voltage V_(C2) (805) will switch to a low value. Transistor M₁₇ (845) is turned off due to the low value of V_(C2) (805) applied at its gate. Transistor M₁₈ (850) is turned on due to the low value of V_(C2) (805) applied at its source. As a result of transistor M₁₈ (850) turning on, the high supply voltage V_(C) (530) is applied to capacitor C₂ (815), charging capacitor C₂ (815) such that capacitor voltage V_(C2) (805) is close to the high supply voltage V_(C) (530). Additionally, both capacitor voltage V_(C3) (860) and the gate drive voltage V_(BG1) (562) will be driven to a voltage (V_(BS)+V_(C))C₃/(C₃+C_(G) _(—) _(SW1)) where C_(G) _(—) _(SW1) is a total gate capacitance of transistor M_(SW1) (563) plus other parasitic capacitances. Consequently, a large value for the gate drive voltage V_(BG1) (562) can be obtained when the capacitance of capacitor C₃ (855) is much larger than C_(G) _(—) _(SW1).

To maximize the gate drive voltage V_(BG1) (562), capacitor C₃ (855) is implemented using both MIM capacitors and MOS capacitors. Although either the high supply voltage V_(C) (530) or the output voltage V_(DD) (535) can be used in place of a bias voltage V_(BS) (862), power efficiency q is found to be slightly higher when the voltage V_(DD) (535) is used. Transistors M₁₄ (830) and M₁₅ (835) are used for reducing the drain-to-gate voltage on transistors M₁₃ (825) and M₁₆ (840), respectively.

Since a deep NWell process can be used, transistor M_(SW1) (563) is connected to the voltage V_(DD) (535) through the transmission gate U₂ (865) to reduce body effect. As a result of reducing the body effect, the on-resistance of the transistor M_(SW1) (563) is also reduced.

A number of embodiments of the invention have been disclosed. Nevertheless, it will be understood that various modifications may be made without departing from the spirit and scope of the present disclosure. For example, the integrator 415 as configured, takes the form of a proportional control circuit that provides a proportional control signal as a function of the difference between the output voltage V_(DD) (102) and the preselected reference voltage. Other types of proportional control circuits are also contemplated by the present invention such as for example, but certainly not limited to, a circuit commonly known as a “bang-bang” controller circuit. Accordingly, other embodiments including circuits configured as proportional control circuits are also within the scope of the following claims. 

What is claimed is:
 1. A method for supplying a regulated output voltage to a load circuit, utilizing a converter circuit configured to switchably couple a time varying input signal to the load circuit, the method comprising the steps of: integrating the difference between the regulated output voltage and a preselected reference voltage to thereby provide a control signal; and switchably coupling the time varying input signal to the load circuit for controlled periods of time when the polarity of the regulated output voltage and the polarity of the time varying signal are the same, said controlled period of time: commencing when the absolute value of the time varying signal becomes greater than the absolute value of the regulated output voltage; and terminating when the absolute value of the time varying input signal becomes greater than the absolute value of the control signal for maintaining the average value of the regulated output voltage equal to the preselected reference voltage.
 2. The method of claim 1 further comprising the step of providing a first voltage being a function of the difference between the time varying input signal and the control signal.
 3. The method of claim 2 further comprising the step of providing a second voltage being a function of the difference between the time varying input signal and the output voltage.
 4. The method of claim 3 further comprising the steps of providing a first drive signal when the second voltage indicates that the absolute value of the time varying signal becomes greater than the absolute value of the output voltage and further providing a second drive signal when the first voltage indicates that the absolute value of the time varying signal becomes greater than the absolute value of the control signal.
 5. The method of claim 4 further comprising the steps of coupling the time varying input signal to the load circuit upon occurrence of the first drive signal and decoupling the time varying input signal from the load circuit upon the occurrence of the second drive signal.
 6. A method of supplying a regulated output voltage to a load circuit, said method utilizing a converter circuit configured to switchably couple a time varying input signal to the load circuit for providing a regulated output voltage having an average value equal to a preselected reference voltage, said method comprising the steps of: providing a proportional control signal as a function of the difference between the regulated output voltage and the preselected reference voltage for providing the proportional control signal; and switchably coupling the time varying input signal to the load circuit for controlled periods of time, said controlled period of time: commencing when the absolute value of the time varying signal becomes greater than the absolute value of the output voltage; and terminating when the absolute value of the time varying input signal becomes greater than the absolute value of the control signal, for thereby maintaining the average value of the output voltage equal to the preselected reference voltage. 